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19-0192; Rev 1; 11/93
High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller
_______________General Description
The MAX746 is a high-efficiency, high-current, step-down DC-DC power-supply controller that drives external N-channel FETs. It provides 93% to 96% efficiency from a 6V supply voltage with load currents ranging from 50mA up to 3A. It uses a pulse-width-modulating (PWM) current-mode control scheme to provide precise output regulation and low output noise. The MAX746's 4V to 15V input voltage range, fixed 5V/adjustable (Dual-ModeTM) output, and adjustable current limit make this device ideal for a wide range of applications. High efficiency is maintained with light loads due to a proprietary automatic pulse-skipping control (Idle-ModeTM) scheme that minimizes switching losses by reducing the switching frequency at light loads. The low 950A quiescent current and ultra-low 1.4A shutdown current further extend battery life. External components are protected by the MAX746's cycleby-cycle current limit. The MAX746 also features a 2V 1.5% reference, a comparator for low-battery detection or level translating, and soft-start and shutdown capability. The MAX747--discussed in a separate data sheet-- functions similarly to the MAX746, but drives P-channel logic level FETs.
____________________________Features
o 93% to 96% Efficiency for 50mA to 3A Output Currents o 4V to 15V Input Voltage Range o Low 950A Supply Current o 1.4A Shutdown Current o Drives External N-Channel FETs o Fixed-Frequency Current-Mode PWM (Heavy Loads) o Idle-Mode PFM (Light Loads) o Cycle-by-Cycle Current Limiting o 2V 1.5% Accurate Reference Output o Adjustable Soft-Start o Undervoltage Lockout o Precision Comparator for Power-Fail or Low-Battery Warning
MAX746
______________Ordering Information
PART MAX746CPE MAX746CSE MAX746C/D MAX746EPE MAX746ESE MAX746MJE TEMP. RANGE 0C to +70C 0C to +70C 0C to +70C -40C to +85C -40C to +85C -55C to +125C PIN-PACKAGE 16 Plastic DIP 16 Narrow SO Dice* 16 Plastic DIP 16 Narrow SO 16 CERDIP
________________________Applications
5V-to-3.3V Green PC Applications Notebook/Laptop Computers Personal Digital Assistants Battery-Operated Equipment Cellular Phones
* Contact factory for dice specifications.
__________Typical Operating Circuit __________________Pin Configuration
INPUT 6V TO 15V
V+
TOP VIEW
AV+ 40m
CP HIGH SHDN ON/OFF LOW-BATTERY DETECTOR INPUT LBI REF SS OUT LBO CC FB AGND GND
LBO 1 LBI 2
16 GND 15 V+ 14 CP
MAX746
CS EXT 39H
OUTPUT 5V
440F
SS 3 REF 4 SHDN 5 FB 6 CC 7 AV+ 8
MAX746
13 HIGH 12 EXT 11 AGND 10 CS 9 OUT
LOW-BATTERY DETECTOR OUTPUT
DIP/SO
TMDual-Mode and Idle-Mode are trademarks of Maxim Integrated Products.
________________________________________________________________ Maxim Integrated Products 1
Call toll free 1-800-998-8800 for free samples or literature.
High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller MAX746
ABSOLUTE MAXIMUM RATINGS
Supply Voltage V+, AV+ to GND ..............................-0.3V to 17V HIGH, EXT to GND....................................................-0.3V to 21V AGND to GND..........................................................-0.3V to 0.3V All Other Pins................................................-0.3V to (V+ + 0.3V) Reference Current (IREF) ....................................................2mA Continuous Power Dissipation (TA = +70C) Plastic DIP (derate 10.53mW/C above +70C) ..........842mW Narrow SO (derate 8.70mW/C above +70C) ............696mW CERDIP (derate 10.00mW/C above +70C) ...............800mW Operating Temperature Ranges: MAX746C_E ........................................................0C to +70C MAX746E_E .....................................................-40C to +85C MAX746MJE ..................................................-55C to +125C Junction Temperatures: MAX746C_E/E_E..........................................................+150C MAX746MJE.................................................................+175C Storage Temperature Range .............................-65C to +160C Lead Temperature (soldering, 10sec) .............................+300C
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(V+ = 10V, ILOAD = 0A, IREF = 0A, TA = TMIN to TMAX, unless otherwise noted.) PARAMETER Input Voltage Output Voltage SYMBOL V+ VOUT V+ = 6V to 15V, 0V < (V+ - CS) < 0.125V, FB = 0V (includes line and load regulation) (V+ - CS) = 0V, external feedback mode V+ = 6V to 15V, FB = 0V V+ = 4V to 15V, external feedback mode 0V < (V+ - CS) < 0.125V Circuit of Figure 1, ILOAD = 0.5A to 2.5A, V+ = 6V VOUT = 5V For dual-mode switchover FB = 2V VREF IREF = 0A IREF = 0A to 100A SS = 0V SS = 2V Operating, V+ = 15V Supply Current (Note 2) ISUPP Operating, V+ = 10V Shutdown mode Oscillator Frequency fOSC MAX746C MAX746E/M 85 80 MAX746C MAX746E/M 0.95 1.4 100 100 20 115 120 A kHz 0.5 100 MAX746C MAX746E/M 1.97 1.96 1 2.00 2.00 9 1.0 500 1.1 1.4 1.7 mA 1.3 94 50 80 40 100 2.03 2.04 20 1.5 MAX746C MAX746E/M CONDITIONS MIN 4 4.85 1.96 1.95 5.08 2.00 2.00 0.05 0.1 2.5 TYP MAX 15 5.25 2.04 2.05 UNITS V V
Feedback Voltage
VFB
V
Line Regulation Load Regulation Efficiency OUT Leakage Current FB Input Logic Low FB Input Leakage Current Reference Voltage Reference Load Regulation Soft-Start Source Current Soft-Start Fault Current (Note 1)
%/V % % A mV nA V mV A A
2
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High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller MAX746
ELECTRICAL CHARACTERISTICS (continued)
(V+ = 10V, ILOAD = 0A, IREF = 0A, TA = TMIN to TMAX, unless otherwise noted.) PARAMETER Maximum Duty Cycle Charge-Pump Output Voltage Current-Sense Amplifier Current-Limit Threshold EXT Output High EXT Output Low EXT Sink Current EXT Source Current Compensation Pin Impedance MAX746C LBI Threshold Voltage LBO Output Voltage Low LBI Input Leakage Current LBO Output Leakage Current SHDN Input Voltage Low SHDN Input Voltage High SHDN Input Leakage Current Note 1: Note 2: VIL VIH SHDN = 10V 2.0 0.1 100 VOL LBI falling MAX746E/M ISINK = 0.5mA LBI = 2.5V V+ = 15V, LBO = 15V, LBI = 2.5V 1.96 2.00 2.04 0.4 100 1 0.4 V nA A V V nA 1.97 VHIGH VLIMIT SYMBOL V+ = 6V IHIGH = 0mA to 10mA V+ - CS VHIGH forced to 15V, IEXT = -1mA VHIGH forced to 15V, IEXT = 1mA VHIGH = 15V, VEXT = 12.5V VHIGH = 15V, VEXT = 2.5V 160 270 24 2.00 2.03 V CONDITIONS MIN 91 V+ + 4 125 VHIGH - 0.1 0.25 TYP 96 V+ + 5 V+ + 6 150 175 MAX UNITS % V mV V V mA mA k
The soft-start fault current is the current sink capability of SS when VREF < 1V or when the device is in shutdown. ISUPP is the supply current drawn by V+, which includes the current drawn by the charge pump. The charge pump doubles the current drawn by HIGH from the V+ input, so ISUPP = IV+ + 2IHIGH.
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3
High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller MAX746
__________________________________________Typical Operating Characteristics
(Circuit of Figure 1a, TA = +25C, unless otherwise noted.)
N0-LOAD SUPPLY CURRENT vs. SUPPLY VOLTAGE
MAX746-01
NO-LOAD SUPPLY CURRENT vs. TEMPERATURE
MAX746-02
CONTINUOUS-CONDUCTION MODE BOUNDARY AND CORRESPONDING PEAK INDUCTOR CURRENT
DISCONTINUOUSCONDUCTION REGION 13 SUPPLY VOLTAGE (V) PEAK INDUCTOR CURRENT
MAX746-09
1.2 NO-LOAD SUPPLY CURRENT (mA)
4 NO-LOAD SUPPLY CURRENT (mA) V+ = 9V VOUT = 5V 3
15
1.1
11
1.0
2 ENTIRE CIRCUIT 1 SCHOTTKY DIODE LEAKAGE EXCLUDED -75 -50 -25 0 25 50 75 100 125
9 7 CONTINUOUSCONDUCTION REGION 0.7 0.9 1.1 1.3 1.5 1.7
0.9
0.8 5 7 9 11 13 SUPPLY VOLTAGE (V) 15
0
5 OUTPUT CURRENT (A)
TEMPERATURE (C)
EFFICIENCY vs. OUTPUT CURRENT
MAX746-07
EFFICIENCY vs. OUTPUT CURRENT
MAX746-08
EFFICIENCY vs. OUTPUT CURRENT
CIRCUIT OF FIGURE 1c VOUT = 5V VIN = 6V EFFICIENCY (%) 90 VIN = 12V
MAX746-06
100
100 CIRCUIT OF FIGURE 1b VOUT = 3.3V V+ = 5V EFFICIENCY (%) 90
100
EFFICIENCY (%)
90
VIN = 6V VIN = 9V VIN = 12V
80
80
80
CIRCUIT OF FIGURE 1a VOUT = 5V 70 0.01 0.1 1 10 OUTPUT CURRENT (A) 70 0.01 0.1 1 10 OUTPUT CURRENT (A) 70 0.01 0.1 1 10 OUTPUT CURRENT (A)
PEAK INDUCTOR CURRENT vs. OUTPUT CURRENT
CIRCUIT OF FIGURE 1a VOUT = 5V 3
MAX746-03
PEAK INDUCTOR CURRENT vs. OUTPUT CURRENT
MAX746-05
PEAK INDUCTOR CURRENT vs. OUTPUT CURRENT
CIRCUIT OF FIGURE 1c VOUT = 5V 1.0 VIN = 12V 0.5
MAX746-03
4 PEAK INDUCTOR CURRENT (A)
4 PEAK INDUCTOR CURRENT (A) CIRCUIT OF FIGURE 1b VOUT = 3.3V V+ = 5V
1.5 PEAK INDUCTOR CURRENT (A)
3
2 VIN = 12V 1 VIN = 9V 0 0.01 0.1 1 10 OUTPUT CURRENT (A) VIN = 6V
2
1
VIN = 6V 0 0.01 0.1 1 10 OUTPUT CURRENT (A) 0 0.01 0.1 1 10 OUTPUT CURRENT (A)
4
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High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller
____________________________Typical Operating Characteristics (continued)
(Circuit of Figure 1a, TA = +25C, unless otherwise noted.)
MAX746
LOAD-TRANSIENT RESPONSE
LOAD-TRANSIENT RESPONSE
LINE-TRANSIENT RESPONSE
10V A
8V A A
B
B
B
200s/div A: LOAD CURRENT, 0.1A TO 1.5A, 1A/div B: VOUT RIPPLE, 50mV/div, AC-COUPLED V+ = 10V
1ms/div A: LOAD CURRENT, 0.1A TO 1.5A, 1A/div B: VOUT RIPPLE, 50mV/div, AC COUPLED V+ = 10V
500ms/div A: V+ = 8V TO 10V, 2V/div B: VOUT RIPPLE, 100mV/div IOUT = 3A
CONTINUOUS-CONDUCTION MODE WAVEFORMS
A
DISCONTINUOUS-CONDUCTION IDLE-MODE WAVEFORMS
MODERATE-LOAD, IDLE-MODE WAVEFORMS
A A
B
B 0V C C
B
C
5s/div A : EXT VOLTAGE, 20V/div B : INDUCTOR CURRENT 1A/div C : VOUT RIPPLE, 50mV/div V+ = 10V, IOUT = 3A
20s/div A: EXT VOLTAGE, 10V/div B: INDUCTOR CURRENT, 500mA/div C: VOUT RIPPLE, 50mV/div, AC-COUPLED V+ = 10V, IOUT = 75mA
20s/div A: EXT VOLTAGE, 10V/div B: INDUCTOR CURRENT, 500mA/div C: VOUT RIPPLE, 50mV/div, AC-COUPLED V+ = 6V, IOUT = 480mA
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5
High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller MAX746
______________________________________________________________Pin Description
PIN 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 NAME LBO LBI SS REF SHDN FB CC AV+ OUT CS AGND EXT HIGH CP V+ GND FUNCTION Low-battery output is an open-drain output that goes low when LBI is less than 2V. Connect to V+ through a pull-up resistor. Leave floating if not used. LBO is disabled in shutdown mode. Input to the low-battery comparator. Tie to V+ or GND if not used. Soft-start limits start-up surge currents. On power-up, it charges the soft-start capacitor, slowly raising the peak current limit to the level set by the sense resistor. 2V reference output can source 100A for external loads. Bypass with 1F. The reference is disabled in shutdown mode. Active-high logic input. In shutdown mode, VOUT = 0V and the supply current is reduced to less than 20A. Connect to GND for normal operation. Feedback input for adjustable-output operation. Connect to GND for fixed 5V output. Use a resistor-divider network to adjust the output voltage (see Setting the Output Voltage section). AC compensation input for the error amplifier. Connect a capacitor between CC and GND for fixed 5V-output operation (see Compensation Capacitor section). Quiet supply voltage for sensitive analog circuitry. Also the noninverting input to the current-sense amplifier. A separate bypass capacitor is not recommended for AV+. Output voltage sense that connects to the internal resistor divider. Bypass with 0.1F to AGND, close to the IC for fixed output operation. Leave unconnected for adjustable-output operation. Inverting input to the current-sense amplifier. Connect the current-sense resistor (RSENSE) from AV+ to CS. Quiet analog ground. Power MOSFET gate-drive output that swings between HIGH and GND. EXT is not protected against short circuits to V+ or AGND. Regulated high-side voltage, 5V above the V+ supply voltage. Charge-pump output that generates a 0V to V+, 50kHz square wave (see Charge Pump section). High-current supply voltage for the charge pump. High-current ground return for the output driver and charge pump.
____________________Getting Started
Figure 1a shows the 5V-output 3A standard application circuit, Figure 1b shows the 3.3V-output 3A standard application circuit, and Figure 1c shows the 5V-output 1.5A standard application circuit. Most applications will be served by these circuits. To learn more about component selection for particular applications, refer to the Design Procedure section. To learn more about the operation of the MAX746, refer to the Detailed Description.
current-mode pulse-width-modulating (PWM) control scheme that results in tight output-voltage regulation, excellent load- and line-transient response, low noise, and high efficiency over a wide range of load currents. Efficiency at light loads is further enhanced by a proprietary idle-mode switching control scheme that skips oscillator cycles in order to reduce switching losses. Other features include undervoltage lockout, shutdown, and a low-battery detection comparator.
_______________Detailed Description
The MAX746 monolithic, CMOS, step-down, switchmode power-supply controller provides high-side drive for external logic-level N-channel FETs. A charge pump generates a voltage 5V above the supply voltage for high-side drive capability. The MAX746 uses a unique
6
Operating Principle
Figure 2 is the MAX746 block diagram. The MAX746 regulates using an inner current-feedback loop and an outer voltage-feedback loop. A slope-compensation scheme stabilizes the current loop; the dominant pole, formed by the output filter capacitor and the load, stabilizes the voltage loop.
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High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller MAX746
V IN 6V TO 15V
C3 0.1F C2 100F D4 1N5817 15 V+ LBI CP HIGH 3 4 6 5 11 SS REF FB SHDN AGND OUT AV+ D2 1N914 * 14 13 8 R SENSE 40m Q1 Si9410DY D3 1N914 * C8 0.1F C9 4.7F
R2 2 R1 C5 0.1F C6 1.0F
MAX746
CS
10
EXT 12 CC 7 C7 2.7nF
N
L1 39H
5V AT 3A
C1 430F
D1 NSQ03A03
9 C4 0.1F 1 R3 100k
LB0 GND 16
* SEE TABLE 2 FOR DIODE SELECTION.
Figure 1a. 5V Standard Application Circuit (15W)
Discontinuous-/ContinuousConduction Modes
The MAX746 is designed to operate in continuous-conduction mode (CCM) but can also operate in discontinuous-conduction mode (DCM), making it ideal for variableload applications. In DCM, the current starts at zero and returns to zero on each cycle. In CCM, the inductor current never returns to zero; it consists of a small AC component superimposed on a DC offset. This results in higher current capability because the AC component in the inductor current waveform is small. It also results in lower output noise, since the inductor does not exhibit the ringing that would occur if the current reached zero (see inductor waveforms in the Typical Operating Characteristics). To transfer equal amounts of energy to the load in one cycle, the peak current level for the discontinuous waveform must be much larger than the peak current for the continuous waveform.
Under these conditions, the inductor must be scaled to the current-sense resistor value. Overcompensation adds a pole to the outer voltage feedback-loop response, degrading loop stability. This may cause voltage-mode pulse-frequency-modulation instead of PWM operation. Undercompensation results in inner current feedback-loop instability, and may cause the inductor current to staircase. Ideal matching between the sense resistor and inductor is not required; it can differ by 30% or more.
Oscillator and EXT Control
The oscillator frequency is nominally 100kHz, and the duty cycle varies from 5% to 96%, depending on the input/output voltage ratio. EXT, which provides the gate drive for the external logic-level N-FET, is switched between HIGH and GND at the switching frequency. EXT is controlled by a unique two-comparator control scheme consisting of a PWM comparator and an idle-mode comparator (Figure 2). The PWM comparator determines the cycle-by-cycle peak current with heavy loads, and the idle-mode comparator sets the light-load peak current. As VOUT begins to drop, EXT goes high and remains high until both comparators trip. With heavy loads, the idle-mode comparator trips first and the PWM control comparator determines the EXT on-time;
7
Slope Compensation
Slope compensation stabilizes the inner current-feedback loop by adding a ramp signal to the current-sense amplifier output. Ideal slope compensation can be achieved by adding a linear ramp, with the same slope as the declining inductor current, to the rising inductor current-sense voltage.
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High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller MAX746
V IN 4.5V TO 6V
C3 0.1F C2 100F D2 D3 D5 1N914 1N914 1N914 D6 1N914 D4 1N5817 C11 1F
R2 2 R1 C5 0.1F C6 1F LBI
15 V+ CP HIGH AV+
14 13 8
C8 0.1F
C9 1F
C10 0.1F
3 4
SS REF
MAX746
CS
10 Q1 Si9410DY
R SENSE 40m
EXT 12 5 11 SHDN AGND FB CC 7 R5 13k (1%) C7 2nF 1 R4 20k (1%)
N
L1 * 22H C4 0.1F
3.3V AT 3A
C3 660F
OUT 9 6
D1 NSQ03A03
R3 100k
LB0 GND 16
*
SUMIDA CDR125 22H SURFACE-MOUNT INDUCTOR
Figure 1b. 3.3V Standard Application Circuit (9.9W)
with light loads, the PWM comparator trips quickly and the idle-mode comparator sets the EXT on-time. Traditional PWM converters continue to switch on every cycle, even when the inductor current is discontinuous due to smaller loads, decreasing light-load efficiency. In contrast, the MAX746's idle-mode comparator increases the switch on-time, allowing more energy to be transferred per cycle. Since fewer cycles are required, the switching frequency is reduced, resulting in minimal switching losses and increased efficiency. The light-load output noise spectrum widens due to the variable switching frequency in idle-mode, but output ripple remains low. Using the Typical Operating Circuit, with a 9V input and a 125mA load current, output ripple is less than 40mV.
quency. When the voltage at HIGH exceeds AV+ by 5V, the charge-pump oscillator is inhibited (Figure 2). When the voltage at HIGH is less than 4.3V below V+, undervoltage lockout occurs. Use the voltage tripler (Figure 3b) when V+ 6V; otherwise, use the voltage doubler (Figure 3a).
Soft-Start and Current Limiting
The MAX746 draws its highest current at power-up. If the power source to the MAX746 cannot provide this initial elevated current, the circuit may not function correctly. For example, after prolonged use the increased series resistance of a battery may prevent it from providing adequate initial surge currents when the MAX746 is brought out of shutdown. Using soft-start (SS) minimizes the possibility of overloading the incoming supply at power-up by gradually increasing the peak current limit. Connect an external capacitor from SS to AGND to reduce the initial peak currents drawn from the supply.
Charge Pump
The MAX746 contains all the control circuitry required to provide a regulated charge-pump voltage 5V above V+ for high-side driving N-channel logic FETs. The charge pump operates with a nominal 50kHz fre-
8
_______________________________________________________________________________________
High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller MAX746
VIN 6V TO 15V
C3 0.1F C2 47F D4 1N5817 15 V+ LBI CP HIGH 3 4 6 5 11 SS REF FB SHDN AGND OUT AV+ D2 1N914 * 14 13 8 R SENSE 75m Q1 Si9410DY D3 1N914 * C8 0.1F C9 4.7F
R2 2 R1 C5 0.1F C6 1F
MAX746
CS
10
EXT 12 CC 7 C7 1nF
N
L1 ** 82H
5V AT 1.5A
C1 220F
D1 NSQ03A03
9 C4 0.1F 1 R3 100k
LB0 GND 16
* SEE TABLE 2 FOR DIODE SELECTION. ** SUMIDA CDR125 SURFACE-MOUNT INDUCTOR.
Figure 1c. 5V Standard Application Circuit (7.5W)
The steady-state SS pin voltage is typically 3.8V. On power-up, SS sources 1A until its voltage reaches 3.8V. The current-limit comparator inhibits EXT switching until the SS voltage reaches 1.8V. The peak current limit is set by: VLIMIT 150mV (typ) IPK = _________ = ___________ RSENSE RSENSE where VLIMIT is the differential voltage across the currentsense amplifier inputs. Figure 4 shows how the SS peak current limit increases as the voltage on SS rises for two RSENSE values.
Shutdown Mode
When SHDN is high, the MAX746 is shut down. In this mode, the internal biasing circuitry (including EXT) is turned off, VOUT drops to 0V, and the supply current drops to 1.4A (20A max). This excludes external component leakage, which may add several microamps to the shutdown supply current for the entire circuit. SHDN is a logic input. Connect SHDN to GND for normal operation.
Low-Battery Detector
The MAX746 provides a low-battery comparator that compares the voltage on LBI to the reference voltage. LBO, an open-drain output, goes low when the LBI voltage is below V REF. Use a resistor-divider network, as shown in the Input Voltage Monitor Circuit (Figure 5), to set the trip voltage (VTRIP) at the desired level. In this circuit, LBO goes low when V+ VTRIP. LBO is high impedance in shutdown mode.
Undervoltage Lockout
Undervoltage lockout inhibits operation of EXT until the charge pump is capable of generating a voltage greater than 4.3V above the supply voltage (Figure 2). When the undervoltage-lockout comparator detects an undervoltage condition, the switching action at EXT is halted.
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9
High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller MAX746
LBO LBI N
EXT
HIGH FROM AV+ 4.3V CHARGE-PUMP CONTROL COMPARATOR T T FLIPFLOP Q UNDERVOLTAGELOCKOUT COMPARATOR 100kHz OSCILLATOR EXT CONTROL
V+
PUMP
LOW-BATTERY COMPARATOR
+2V REFERENCE
5V
REF
OUT
CC
SHDN
FB
ERROR AMPLIFIER
PWM COMPARATOR
DUAL-MODE COMPARATOR 100mV AV+ CURRENT-SENSE AMPLIFIER
CS
SLOPECOMPENSATION RAMP
VRAMP
50mV
LIGHT-LOAD COMPARATOR
SS
SOFT-START CIRCUITRY
CURRENT-LIMIT COMPARATOR
AGND
GND
Figure 2. Block Diagram
10
______________________________________________________________________________________
High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller MAX746
3a. CHARGE-PUMP VOLTAGE DOUBLER 3 V IN 15 V+ D2 1N914 C8 0.1F D3 1N914 D4 1N5817 C9 1F PEAK CURRENT LIMIT (A) 2 V+ - VCS = 150mV
PEAK CURRENT LIMIT vs. SOFT-START VOLTAGE
MAX746-FG03
RSENSE = 50m
MAX746
T FLIPT FLOP Q CLK 100kHz OSCILLATOR 5V AV+ GND 16 V IN D3 D2 1N914 1N914 C8 0.1F CP 14 13 3b. CHARGE-PUMP VOLTAGE TRIPLER D4 1N5817 D5 1N914 C9 1F D6 1N914 C10 0.1F C11 1F CP 14 HIGH 13
1
RSENSE = 100m 0 0 1 2 3 SOFT-START VOLTAGE (V) 4
Figure 4. Peak Current Limit vs. Soft-Start Voltage
VIN 15 V+ ...TO VOUT OR VIN R3 100k LBO 1 LOW-BATTERY OUTPUT GND 16 R2 = R1 R2 2
15 V+
MAX746
LBI
MAX746
GND 16
HIGH
R1
VREF = 2.0V
( VTRIP -1 ) VREF
Figure 3. Charge-Pump Configurations
Figure 5. Input Voltage Monitor Circuit
__________________Design Procedure
Setting the Output Voltage
The MAX746's dual-mode output voltage can be set to 5V by grounding FB, or it can be adjusted from 2V to 14V using external resistors R4 and R5 configured as shown in Figure 6. Select feedback resistor R4 in the 10k to 60k range. R5 is given by: VOUT R5 = (R4) _______ - 1 2V The MAX746 is designed to use either internal or external feedback mode, but should not be toggled between
the two modes while operating. If two different output voltages are required, use external feedback mode with a resistor network similar to the 3.3V/5V adjustable output circuit shown in Figure 7.
(
)
Selecting RSENSE To select the sense-resistor value (R SENSE ), first approximate the peak current assuming I PK is (1.1) (ILOAD), where ILOAD is the maximum load current. Once all component values have been determined, the actual peak current is given by:
VOUT IPK = ILOAD + ___________ (2L) (fOSC)
(
OUT )(1- _______) VIN
V
______________________________________________________________________________________
11
High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller MAX746
VIN 15 V+ FB 6 C7* R4 R5 VOUT
EXT 12 N L D1 R5 26.1k (1%) FB 6 OUT 9 R4a 17.4k (1%) 5V/3.3V N C7 V OUT C1
MAX746
MAX746
OUT GND 16 9 R4 = 10k TO 60k VOUT R5 = R4 -1 VREF VREF = 2.0V NOMINAL
SELECT WITH FET OFF: VOUT = VREF SELECT WITH FET OFF: VOUT = VREF
(
)
R5 ( R4a +1 )
R4b 22.6k (1%)
* SEE COMPENSATION CAPACITOR SECTION.
Figure 6. Adjustable Output Circuit
( R4a R5R4b +1 ) +
VREF = 2.0V NOMINAL
Figure 7. 3.3V/5V Ajustable Output Circuit
Next, determine the value of RSENSE such that: VLIMIT(min) 125mV RSENSE = _____________ = ________ IPK IPK For example, to obtain 5V at 3A, I PK = 3.3A and RSENSE = 125mV/3.3A = 38m. The sense resistor should have a power rating greater than (I PK2) (RSENSE) with an adequate safety margin. With a 3A load current, IPK = 3.3A and RSENSE = 38m. The power dissipated by the resistor (assuming an 80% duty cycle) is 331mW. Metal-film resistors are recommended. Do not use wire-wound resistors because their inductance will adversely affect circuit operation. The duty cycle (for continuous conduction) is determined from the following equation: VOUT + VDIODE Duty Cycle (%) = _____________________ x 100% V+ - VSW + VDIODE where V SW is the voltage drop across the external N-FET and sense resistor. VSW can be approximated as [ILOAD x (rDS(ON) + RSENSE)].
where VRAMP(max) is the 50mV peak value of the slopecompensation linear ramp signal. Although 38H is the calculated value, the component used may have a tolerance of 30% or more. Inductors with molypermalloy powder (MPP), Kool M, or ferrite are recommended. Inexpensive iron-powder core inductors are not suitable, due to their increased core losses, especially at switching frequencies in the 100kHz range. MPP and Kool M cores have low permeability, allowing larger currents. For highest efficiency, use a coil with low DC resistance. To minimize radiated noise, use a toroid, a pot core, or a shielded coil. It is customary to select an inductor with a saturation rating that exceeds the peak current set by R SENSE, but inductors are often specified very conservatively. If the inductor's core losses do not cause excessive temperature rise (inductor wire insulation is usually rated for +125C) and the associated efficiency losses are minimal, inductors with lower current ratings are acceptable. In the 3.3V Standard Application Circuit (Figure 1b), the inductor selected has a 2.2A current rating even though the peak current is 3.3A. This inductor was selected for two reasons: it is the highest-rated readily available surface-mount inductor of its size, and lab tests have verified that the core-loss increase is minimal. With a 3A load current, the inductor current does not begin showing significant losses due to saturation until the supply voltage increases to 10V (the maximum supply for this circuit is 6V).
Inductor Selection
Once the sense-resistor value is determined, calculate the inductor value (L) using the following equation. The correct inductor value ensures proper slope compensation. Continuing from the equations above:
(RSENSE) (VOUT) L = ______________________ (VRAMP(max)) (fOSC) (50mV) (100kHz) (38m) (5V) = _____________________ = 38H
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High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller
External Logic-Level N-FET Selection
To ensure the external N-FET is turned on hard, use logic-level or low-threshold N-FETs. Three important parameters to note when selecting the N-FET are the total gate charge (Qg), on resistance (rDS(ON)), and reverse transfer capacitance (CRSS). Qg includes all capacitances associated with charging the gate. Use the typical Qg value for best results; the maximum value is usually grossly overspecified, since it is a guaranteed limit and not the measured value. The typical total gate charge should be 50nC or less; with larger numbers, EXT may not be able to adequately drive the gate. EXT sink/source capability (IEXT) is typically 210mA. The two most significant losses contributing to the N-FET's power dissipation are I2R losses and switching losses. CCM power dissipation (PD), is approximated by: PD = (Duty Cycle) (IPK2) (rDS(ON)) + To ensure stability, the minimum capacitance and maximum ESR values are:
MAX746
(5) (VREF) C1(min) > ______________________________ (2) (GBW) (VOUT) (RSENSE)
and,
(VOUT ) (RSENSE) ESRC1 < ___________________ (VREF)
where GBW = the loop gain-bandwidth product, 15kHz. Sprague 595D surface-mount solid tantalum capacitors and Sanyo OS-CON through-hole capacitors are recommended due to their extremely low ESR. OS-CON capacitors are particularly useful at low temperatures. For best results when using other capacitors, increase the output filter capacitor's size or use capacitors in parallel to reduce the ESR. Bypass OUT with a 0.1F (C4) capacitor to GND when using a fixed 5V output (Figures 1a and 1c). With adjustable-output operation, place C4 between the output voltage and AGND as close to the IC as possible (Figure 1b). The circuit load-step response is improved by using a larger output filter capacitor or by placing a low-cost bulk capacitor in parallel with the required low-ESR output filter capacitor. The output voltage sag under a load step (ISTEP) is approximated by:
STEP VSAG = _____________________________________ (2) (C1) (VIN(MIN) (DMAX - VOUT)
(V+ ) (CRSS) (IPK) (fOSC) __________________________
2
(IEXT)
where the duty cycle is approximately V OUT /V+, fOSC = 100kHz, and rDS(ON) and CRSS are given in the data sheet of the chosen N-FET. In the equation, rDS(ON) is assumed constant, but is actually a function of temperature. The equation given does not account for losses incurred by charging and discharging the gate capacitance, because that energy is dissipated by the gate-drive circuitry, not the N-FET. The Standard Application Circuits (Figure 1) use an 8-pin, Si9410DY, surface-mount N-FET that has 0.05 on resistance with a 4.5V V GS. Optimum efficiency is obtained when the voltage at the source swings between the supply rails (within a few hundred millivolts).
(I
2)
(L)
Diode Selection
The MAX746's high switching frequency demands a high-speed rectifier. Schottky diodes are recommended. Ensure that the Schottky diode average current rating exceeds the maximum load current.
where DMAX is the maximum duty cycle (91% worst case). The equation assumes an input/output voltage differential of 2V or more. Table 1 gives measured values of output voltage sag with a 30mA to 3A load step for various input voltages and output filter capacitors. Refer also to the AC Stability with Low Input/Output Differentials section.
Capacitor Selection
Output Filter Capacitor The output filter capacitor C1 should have a low effective series resistance (ESR), and its capacitance should remain fairly constant over temperature. This is especially true when in CCM, since the output filter capacitor and the load form the dominant pole that stabilizes the voltage loop.
Input Bypass Capacitor The input bypass capacitor C2 reduces peak currents drawn from the voltage source, and also reduces the amount of noise at the voltage source caused by the MAX746's fast switching action (this is especially important when other circuitry is operated from the same source). The input capacitor ripple current rating must exceed the RMS input ripple current.
IRMS = RMS AC input current = ILOAD
(V ) (V - V ) (_______________________) V
OUT IN OUT IN 13
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High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller MAX746
Table 1. Measured Output Voltage Sag with 30mA to 3A Load Step*
OUTPUT FILTER CAPACITOR C1 (mF) 440 660 880 OUTPUT VOLTAGE SAG (mV) FOR VARIOUS INPUT VOLTAGES
VIN =6V VIN=6.5V VIN=7V VIN=9V VIN =10V
Table 2. Charge-Pump Configuration
V+ V+ 6V 6V < V+ < 6.5V* V+ 6.5V* CHARGE-PUMP CONFIGURATION Voltage tripler with 1N914 diodes for D2, D3, D5, and D6 Voltage doubler with 1N5817 Schottky diodes for D2 and D3 Voltage doubler with 1N914 diodes for D2 and D3
400 260 200
250 190 100
210 160 90
140 70 40
90 50 25
*Circuit of Figure 1a.
For load currents up to 3A, 100F (C2) in parallel with 0.1F (C3) is adequate. Smaller bypass capacitors may also be acceptable for lighter loads. The input voltage source impedance determines the size of the capacitor required at the V+ input. As with the output filter capacitor, a low-ESR capacitor (Sanyo OS-CON, Sprague 595D or equivalent) is recommended for input bypassing.
* When using the voltage-doubler circuit over the military
temperature range, increase the 6.5V limit to 7V.
voltage and load current. With a 3A load current, a 10V input voltage, and a 0.1F soft-start capacitor, it typically takes 240ms for the MAX746 to power up. A 0.47F soft-start capacitor increases the start-up time to approximately 2.3sec. Bypass REF with a 1F capacitor (C6).
Charge-Pump Capacitors Figure 3a shows the charge-pump doubler circuit configured with a 0.1F charge-pump capacitor C8 and a 1.0F reservoir capacitor C9. The ratio of the capacitors, along with the input voltage, determines the amount of ripple on HIGH. If the input supply range exceeds 12V, increase C9 to 4.7F to reduce the charge-pump ripple. C9 should be 10F for less. Figure 3b shows the charge-pump tripler circuit. Refer to Table 2 to determine the proper charge-pump configuration (which is based on the minimum expected supply voltage at V+). Some interaction occurs between the switch oscillator and the charge-pump oscillator. This interaction modulates the inductor-current waveform, but has negligible impact on the output. Soft-Start and Reference Capacitors Soft-start provides a ramp to the full current limit. A typical value for the soft-start capacitor (C5) is 0.1F, which provides a 380ms soft-start time. Use values in the 0.001F to 1F range. The nominal time for C5 to reach its steady-state value is given by:
tSS (sec) = (C5) (3.8 x 106) Note that tSS does NOT equal the time it takes for the MAX746 to power-up, although it does affect the startup time. The start-up time is also a function of the input
Compensation Capacitor With a fixed 5V output, connect a compensation capacitor (C7) between CC and AGND to optimize transient response. Appropriate compensation is determined by the size and ESR of the output filter capacitor (C1), and by the load current. In the standard 5V application circuit, 2.7nF is appropriate for load currents up to 3A; for lighter loads, C7's value can be reduced. If 2.7nF does not compensate adequately, use the following equations to determine C7. For fixed 5V-output operation:
(C1) (ESRC1) C7 = _____________ 12k
For adjustable-output operation, FB becomes the compensation input pin, and CC and OUT are left unconnected. Connect C7 between FB and GND in parallel with R4 (Figure 6). C7 is determined by: C7 = ___________________ R4 R5 For example, with a fixed 5V output with C1 = 470F and an ESRC1 of 0.04 (at a frequency of 100kHz):
(2) (C1) (ESRC1)
(C1) (ESRC1) C7 = _____________ = 1560pF 12k
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High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller
Increasing C7 by up to 50% enhances outer-loop stability by adding stability to the inductor current waveform. But increasing C7 too much causes FB's response time to decrease (due to the larger RC time constant caused by the feedback resistors and the compensation capacitor), which reduces load-transient stability.
MAX746
VIN V+ AV+ KELVIN SENSE CONNECTION
RSENSE
Setting the Low-Battery Detector Voltage
Select R1 between 10k and 1M. Determine R2 using the following equation: R2 = R1
MAX746
CS
EXT
N L1 VOUT
(
(VTRIP - VREF) ________________ VREF
)
where VREF is typically 2.0V. Connect a pull-up resistor (e.g., 100k) between LBO and VOUT (Figure 5).
Using a Second Supply in Place of the Charge Pump
If a secondary power supply (a minimum of 5V above the main supply) is available, it can be substituted for the charge-pump high-side supply. In this case, bypass HIGH with a 1F capacitor and leave CP unconnected. Since this secondary supply voltage is applied to the gate, V GS must not exceed the gate-source breakdown voltage of the external N-FET. Also, the voltage at HIGH must not exceed 20V. If a secondary supply is used, the shutdown function cannot be used because HIGH is internally tied to V+ in shutdown mode. In this case, SHDN must be tied low. With the main supply off and HIGH at 12V, HIGH will typically sink 130A.
Figure 8. Kelvin Connection for Current-Sense Amplifier
tor, any noise at the CS input will also appear at the AV+ input, and will be interpreted by the currentsense amplifier as a common-mode signal . A separate AV+ capacitor causes the noise to appear on only one input, and this differential noise will be amplified, adversely affecting circuit operation.
Additional Notes
When probing the MAX746 circuit, avoid shorting V+ to GND (the two pins are adjacent) as this may cause the IC to malfunction because of large ground currents. Because of its fast switching and high drivecapability requirements, EXT is a low-impedance point that is not short-circuit protected. Therefore, do not short EXT to any node (including AGND and V+, which are adjacent to EXT). Similarly, CC (or FB in adjustable-output operation) is a sensitive input that should not be shorted to any node. Avoid shorting CC when probing the circuit, as this may damage the device. The MAX746 may continue to operate with AV+ disconnected, but erratic switching waveforms will appear at EXT.
Layout Considerations
Because high current levels and fast switching waveforms radiate noise, proper PC board layout is essential. Use a ground plane, and minimize ground noise by connecting GND, the anode of the steering Schottky diode, the input bypass-capacitor ground lead, and the output filter capacitor ground lead to a single point (star ground configuration). Also minimize lead lengths to reduce stray capacitance, trace resistance, and radiated noise. Place bypass capacitor C3 as close to V+ and GND as possible. AV+ and CS are the inputs to the differential-input current-sense amplifier. Use a Kelvin connection across the sense resistor, as shown in Figure 8. Although AV+ also functions as the supply voltage for sensitive analog circuitry, a separate AV+ bypass capacitor should not be used. By not using a capaci-
Switching Waveforms There is a region between CCM and DCM where the inductor current operates in both modes, as shown in the Idle-Mode Moderate Current EXT waveform in the Typical Operating Characteristics. As the output voltage varies, it is fed back into CC and the duty cycle adjusts to compensate for this change. The switch is considered off when VEXT is less than
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15
High-Efficiency, PWM, Step-Down, N-Channel DC-DC Controller
or equal to the N-FET's V GS threshold voltage. Once the switch is off, the voltage at EXT is pulled to GND and the N-FET source voltage is a Schottky diode drop below GND. However, this is not always the case in the "in-between" mode, due to the changing duty cycle inherent with DCM. When the device is at maximum duty cycle, EXT turns off at V GS, but the switch sometimes turns on again after the minimum off-time before EXT can be pulled to GND. This results in short spikes, which can be seen on the EXT waveform in the Typical Operating Characteristics.
AC Stability with Low Input/Output Differentials At low input/output differentials, the inductor current cannot slew quickly enough to respond to load changes, so the output filter capacitor must hold up the voltage as the load transient is applied. In Figure 1a's circuit, for V+ = 6V, increase the output filter capacitor to 900F (Sprague 595D low-ESR capacitors) to obtain a transient response less than 250mV with a load step from 0.1A to 3A. As V+ increases, the inductor current slews faster, so the size of the output filter capacitor can be reduced (see Table 1).
MAX746
Table 3. Component Suppliers
SUPPLIER INDUCTORS Coiltronics Gowanda Sumida USA Sumida Japan CAPACITORS Kemet Matsuo Nichicon Sprague Sanyo USA Sanyo Japan United Chemi-Con DIODES Motorola Nihon USA Nihon Japan POWER TRANSISTORS Harris International Rectifier Siliconix RESISTORS IRC (512) 992-7900 (512) 992-3377 (407) 724-3739 (213) 772-2000 (408) 988-8000 (407) 724-3937 (213) 772-9028 (408) 727-5414 (800) 521-6274 (805) 867-2555 81-3-3494-7411 (805) 867-2698 81-3-3494-7414 (803) 963-6300 (714) 969-2491 (708) 843-7500 (603) 224-1961 (619) 661-6322 81-3-3837-6242 (714) 255-9500 (714) 255-9400 (803) 963-6322 (714) 960-6492 (708) 843-2798 (603) 224-1430 (305) 781-8900 (716) 532-2234 (708) 956-0666 81-3-3607-511 (305) 782-4163 (716) 532-2702 (708) 956-0702 81-3-3607-5428 PHONE FAX
___________________Chip Topography
LBI LBO GND V+
SS
CP
HIGH REF SHDN EXT 0.130" (3.30mm)
AGND
FB CC
AV+ OUT 0.080" (2.03mm)
CS
TRANSISTOR COUNT: 508; SUBSTRATE CONNECTED TO HIGH.
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
16 __________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 (408) 737-7600 (c) 1993 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.

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